Radio receiver

ABSTRACT

A despreader  106  ( 107 ) provides despread processing to baseband signals based on timing from a searcher  105  to obtain a baseband signal of path  1  (path  2 ). A complex correlation section  115  ( 116 ) provides complex correlation processing to the baseband signals from the despreader  106 ( 107 ) (baseband signals, which are not yet multiplied by an inverse characteristic of a channel estimation value from a channel estimating section  108 ). A path combining section  117  combines the signals subjected to complex correlation processing by the complex correlation sections  115  and  116.  A phase estimating section  118  estimates a phase rotation amount caused by a frequency offset using the signal combined by the path combining section  117.

TECHNICAL FIELD

The present invention relates to a radio receiving apparatus and moreparticularly to a radio receiving apparatus that compensates for afrequency offset.

BACKGROUND ART

Conventionally, a receiver-side apparatus performs processing(hereinafter referred to as “frequency offset compensation) forcompensating for a carrier frequency drift between a transmitter-sideapparatus (for example, base station apparatus) and a receiver-sideapparatus (for example, communication terminal apparatus).

The following will explain the radio receiving apparatus that performs aconventional frequency offset compensation with reference to FIG. 1,FIG. 2, and FIG. 3. FIG. 1 is a schematic view showing a state of aknown symbol transmitted to the radio receiving apparatus that performsa frequency offset compensation. FIG. 2 is a block diagram showing aconfiguration of the radio receiving apparatus that performs aconventional frequency offset compensation. FIG. 3 is a schematic viewshowing a timing state of a known symbol of a path received by the radioreceiving apparatus that performs a conventional frequency offsetcompensation.

The transmitter-side apparatus (not shown) transmits a signal includinga known symbol 11 spread with Code A and a known symbol 12 spread withCode B. In this case, it is assumed that a length of Code A and a lengthof Code B are set to tCA and tCB, respectively, and that a distancebetween the known symbol 11 and known symbol 12 is set to tgap.

The signal transmitted from the transmitter-side apparatus is receivedvia an antenna 21 by a radio receiving apparatus shown in FIG. 2. InFIG. 2, a signal received (received signal) by the antenna 21 isconverted into a baseband signal from a carrier frequency signal by areception RF section 22. At this time, a local signal sent from acrystal oscillator 38 (to be described later) is used at the receptionRF section 22. An in-phase component (I-ch) of the baseband signal and aquadrature phase component (Q-ch) thereof are output to an A/D converter23 and an A/D converter 24 from the reception RF section 22,respectively.

The baseband signal with I-ch and the baseband signal with Q-ch areconverted into digital signals by the A/converter 23 and A/D converter24, respectively. The baseband signal with I-ch and the baseband signalwith Q-ch, which are converted into digital signals, are output to asearcher 25, a despreader 26, and a despreader 27.

The searcher 25 examines the correlation between the baseband signalconverted to the digital signal and Code A, which is the known code, todetect a code timing (namely, timing of each path) with which power of acorrelation value reaches a maximum value as illustrated in FIG. 3. Thesearcher 25 also detects timing of Code B using the detected codetiming. For example, if a timing difference between path 1 of Code A andpath 2 thereof is set to tp, timing of code B of path 1 becomes tA+tgap,and timing of Code B of path 2 becomes tA+Tgap+tp. Thus, timing of CodeB is also calculated based on the detected timing of Code A. In thisway, despread timing at the despreaders 26 and 27, pilot timing at achannel estimating section 28 and path timing at a RAKE combiningsection 29 are calculated by the searcher 25.

Timing of Code A and Code B of path 1 is output to the despreader 26from the searcher 25, and timing of Code A and Code B of path 2 isoutput to the despreader 27 from the searcher 25. Timing of Code A andCode B of path 1, and timing of Code A and Code B of path 2 are outputto the channel estimating section 28 from the searcher 25. Moreover,timing of path 1 and timing of path 2 are output to the RAKE combiningsection 29 from the searcher 25.

The despreader 26 provides despread processing using Code A and Code Bto the baseband signal with I-ch based on timing of Code A and Code B ofpath 1 from the searcher 25. Similarly, the despreader 26 providesdespread processing to the baseband signal with Q-ch using Code A andCode B based on timing of Code A and Code B of path 1 from the searcher25, respectively. Moreover, the despreader 26 provides despreadprocessing to the baseband signals with I-ch and Q-ch using apredetermined spreading code (spreading code assigned to the presentradio receiving apparatus). The baseband signals with I-ch and Q-chsubjected to despread processing are output to the channel estimatingsection 28 and RAKE combining section 29.

The despreader 27 provides despread processing using Code A and Code Bto the baseband signal with I-ch based on timing of Code A and Code B ofpath 2 from the searcher 25. Similarly, the despreader 27 providesdespread processing to the baseband signal with Q-ch using Code A andCode B based on timing of Code A and Code B of path 2 from the searcher25, respectively. Moreover, the despreader 27 provides despreadprocessing to the baseband signals with I-ch and Q-ch using apredetermined spreading code. The baseband signals with I-ch and Q-chsubjected to despread processing are output to the channel estimatingsection 28 and RAKE combining section 29.

The channel estimating section 28 extracts a signal, which correspondsto the known symbol 11 and known symbol 12, from among baseband signalswith I-ch and Q-ch subjected to spread processing from the despreader 26based on timing of Code A and Code B of path 1 from the searcher 25. Achannel estimation value of path 1 is calculated using this extractedsignal. Likewise, the channel estimating section 28 extracts a signal,which corresponds to the known symbol 11 and known symbol 12, from amongbaseband signals with I-ch and Q-ch subjected to spread processing fromthe despreader 27 based on timing of Code A and Code B of path 2 fromthe searcher 25. A channel estimation value of path 2 is calculatedusing this extracted signal. The channel estimation values of path 1 andpath 2 calculated by the channel estimating section 28 are output to theRAKE combining section 29.

The RAKE combining section 29 multiplies the baseband signal with I-chand Q-ch subjected to despread processing from the despreader 26 by aninverse characteristic of the channel estimation value of path 1 fromthe channel estimating section 28. The RAKE combining section 29multiplies the baseband signal with I-ch and Q-ch subjected to despreadprocessing from the despreader 27 by an inverse characteristic of thechannel estimation value of path 2 from the channel estimating section28. Moreover, the RAKE combining section 29 RAKE combines the despreadbaseband signal with I-ch and Q-ch of path 1 multiplied by the inversecharacteristic of channel estimation value with the despread basebandsignal with I-ch and Q-ch of path 2 multiplied by the inversecharacteristic of channel estimation value based on timing of path 1 andpath 2 from the searcher 25.

The baseband signal with I-ch and Q-ch subjected to RAKE combining isoutput to a modulating section 30. The modulating section 30 providesdemodulation processing to the baseband signal with I-ch and Q-chsubjected to RAKE combining, whereby obtaining received data.

The baseband signal with I-ch subjected to RAKE combining is output to acomplex correlation calculating section 33. Also, after the basebandsignal with I-ch subjected to RAKE combining is delayed by tAB(=tCA/2+tgap+tCB/2; see FIG. 1) by a delay section 31, and the resultantis output to the complex correlation calculating section 33. Similarly,the baseband signal with Q-ch subjected to RAKE combining is output to acomplex correlation calculating section 33. Likewise, after the basebandsignal with Q-ch subjected to RAKE combining is delayed by tAB by adelay section 32, and the resultant is output to the complex correlationcalculating section 33.

The complex correlation calculating section 33 performs complexcorrelation processing using the baseband signal with I-ch subjected toRAKE combining from the RAKE combining 29 and the baseband signal withI-ch subjected to RAKE combining and delayed by tAB from the delaysection 31. Moreover, the complex correlation calculating section 33performs complex correlation processing using the baseband signal withQ-ch subjected to RAKE combining from the RAKE combining 29 and thebaseband signal with Q-ch subjected to RAKE combining and delayed by tABfrom the delay section 32. The signals with I-ch and Q-ch subjected tocomplex correlation processing are output to a phase estimating section34.

The phase estimating section 34 calculates a phase rotation amount perunit time using the signals with I-ch and Q-ch, which are subjected tocomplex correlation processing and which are sent from the complexcorrelation calculating section 33. A smoothing section 35 calculates afrequency offset using the phase rotation amount calculated by the phasecalculating section 34. The calculated frequency offset is output to acontrol voltage converting section 36.

The voltage converting section 36 converts the calculated frequencyoffset into a control voltage to the crystal oscillator 38. This controlvoltage is converted into an analog signal by a D/A converter 37, andthe resultant is output to the crystal oscillator 38. In this way, thefrequency of local signal is controlled at the crystal oscillator 38.The frequency offset compensation is thus carried out.

However, the conventional radio receiving apparatus that performs thefrequency offset compensation has the following problem. Namely, in theconventional radio receiving apparatus that performs the frequencyoffset compensation, since the phase rotation amount is estimated usingthe baseband signals subjected to RAKE combining, there is a problem inwhich accuracy of phase rotation amount to be estimated is decreasedparticularly when Dopplar frequency caused by high-speed moving becomeshigh.

For example, as illustrated in FIG. 4, a channel estimation value iscalculated using a known symbol placed at the central portion of theslot. In a case where the use of this channel estimation value is sharedin the slot, accuracy of the channel estimation value deteriorates withdistance from the channel estimation segment, so that accuracy of thebaseband signals subjected to RAKE combining deteriorates. As a result,accuracy of phase rotation amount to be estimated is decreased. In otherwords, accuracy of phase rotation amount to be estimated depends onaccuracy of channel estimation using the baseband signals subjected toRAKE combining.

The factors that decrease accuracy of the phase rotation amount to beestimated will be explained with reference to FIGS. 5A, 5B, 5C, 5D, 5E,5F, 6A, 6B, and 6C.

FIG. 5A is a schematic view showing a state of the phase rotation amountof the baseband signal obtained by despread processing using Code A ofpath 1. FIG. 5B is a schematic view showing a state of the phaserotation amount of the baseband signal obtained by despread processingusing Code B of path 1.

As illustrated in FIGS. 5A and 5B, since channel estimation of path 1 iscarried out on a path-by-path basis by the channel estimating section28, a channel estimation value obtained using Code A and a channelestimation value obtained using Code B are substantially the same aseach other (Δθ1ch).

The baseband signal obtained by despread processing using Code A(hereinafter simply referred to as “baseband signal of Code A”) rotatesagainst a transmit signal by a phase variation (Δθ1fad) due to fading.The baseband signal obtained by despread processing using Code B(hereinafter simply referred to as “baseband signal of Code B”) rotatesagainst the baseband signal of Code A by (66 θAB).

FIG. 5C is a schematic view showing a state of a phase rotation amountof the baseband signal obtained by despread processing using Code A ofpath 2. FIG. 5D is a schematic view showing a state of a phase rotationamount of the baseband signal obtained by despread processing using CodeB of path 2.

Because of a difference between path 1 and path 2 in the propagationpath, the baseband signal of Code A rotates against the baseband signalof Code A of path 1 by a phase rotation amount (Δθp) and a phasevariation (Δθ2fad) due to fading. In addition, the phase rotation amount(Δθp) is a phase rotation amount corresponding to a time difference (tp)between path 1 and path 2. The baseband signal of Code B further rotatesagainst the baseband signal of Code A of path 1 by ΔθAB.

Next, attention will be paid on the baseband signal subjected to RAKEcombining by the RAKE combining section 29. FIG. 5E is a schematic viewshowing a state of a phase rotation amount of the baseband signal ofCode A subjected to RAKE combining. FIG. 5F is a schematic view showinga state of a phase rotation amount of the baseband signal of Code Bsubjected to RAKE combining.

As illustrated in FIG. 5E, the baseband signal subjected to RAKEcombining of Code A (namely, the baseband signal, which is obtained byRAKE combining the baseband signal of Code A of path 1 and the basebandsignal of Code A of path 2) becomes a signal including a channelestimation error (Δθch. errA).

Likewise, as illustrated in FIG. 5F, the baseband signal subjected toRAKE combining of Code B (namely, the baseband signal, which is obtainedby RAKE combining the baseband signal of Code B of path 1 and thebaseband signal of Code B of path 2) becomes a signal including achannel estimation error (Δθch. errB) and a phase rotation amount (ΔθAB)due to a frequency offset to be calculated.

FIG. 6A is a schematic view showing a state of the channel estimationerror of the baseband signal of Code A subjected to RAKE combining. FIG.6B is a schematic view showing a state of the channel estimation errorof the baseband signal of Code B subjected to RAKE combining. FIG. 6C isa schematic view showing a state of a signal to be subjected to complexcorrelation processing at the conventional radio receiving apparatusthat performs frequency offset compensation.

In other words, as illustrated in FIGS. 6A, 6B, and 6C, since complexcorrelation processing is performed using the baseband signals of Code Aand Code B each which includes the channel estimation error and which issubjected to RAKE combining, the signal obtained by this complexcorrelation processing includes the channel estimation error.Accordingly, the finally obtained phase rotation amount due to thefrequency offset includes a channel estimation error, that is, an errorcorresponding to (Δθch. errA+Δ θ ch. errB). As a result, at thehigh-speed moving time at which channel estimation accuracydeteriorates, the estimation error of the phase rotation amount due toparticularly the frequency offset deteriorates, resulting in a declinein the quality of a demodulated signal.

DISCLOSURE OF INVENTION

The present invention has been made with consideration given to theaforementioned problem, and it is an object of the present invention isto provide a radio receiving apparatus that correctly estimates a phaserotation amount due to a frequency offset even at a high-speed movingtime.

The above object is attained by providing complex correlation processingto a received signal (baseband signal), which is not yet multiplied byan inverse characteristic of a channel estimation value estimated by useof the received signal, to estimate the phase rotation amount in thereceived signal.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a schematic view showing a state of a known symbol transmittedto a radio receiving apparatus that performs frequency offsetcompensation;

FIG. 2 is a block diagram showing a configuration of a conventionalradio receiving apparatus that performs frequency offset compensation;

FIG. 3 is a schematic view showing a timing state of a known symbol in apath received by a conventional radio receiving apparatus that performsfrequency offset compensation;

FIG. 4 is a schematic view showing a state of channel estimationaccuracy due to a conventional radio receiving apparatus that performsfrequency offset compensation;

FIG. 5A is a schematic view showing a state of a phase rotation amountof a baseband signal obtained by despread processing using Code A ofpath 1;

FIG. 5B is a schematic view showing a state of a phase rotation amountof a baseband signal obtained by despread processing using Code B ofpath 1;

FIG. 5C is a schematic view showing a state of a phase rotation amountof a baseband signal obtained by despread processing using Code A ofpath 2;

FIG. 5D is a schematic view showing a state of a phase rotation amountof a baseband signal obtained by despread processing using Code B ofpath 2;

FIG. 5E is a schematic view showing a state of a phase rotation amountof a baseband signal of Code A subjected to RAKE combining;

FIG. 5F is a schematic view showing a state of a phase rotation amountof a baseband signal of Code B subjected to RAKE combining;

FIG. 6A is a schematic view showing a state of a channel estimationerror of a baseband signal of Code A subjected to RAKE combining;

FIG. 6B is a schematic view showing a state of a channel estimationerror of a baseband signal of Code B subjected to RAKE combining;

FIG. 6C is a schematic view showing a state of a signal to be subjectedto complex correlation processing at the conventional radio receivingapparatus that performs frequency offset compensation;

FIG. 7 is a block diagram showing a configuration of a radio receivingapparatus according to Embodiment 1 of the present invention;

FIG. 8 is a schematic view showing a state of a known symbol in a pathto be received by the radio receiving apparatus according to Embodiment1 of the present invention;

FIG. 9A is a schematic view showing a state of a phase rotation amountof a baseband signal obtained by despread processing using Code A ofpath 1;

FIG. 9B is a schematic view showing a state of a phase rotation amountof a baseband signal obtained by despread processing using Code A ofpath 2;

FIG. 9C is a schematic view showing a state of a phase rotation amountof a baseband signal obtained by despread processing using Code B ofpath 1;

FIG. 9D is a schematic view showing a state of a phase rotation amountof a baseband signal obtained by despread processing using Code B ofpath 2;

FIG. 9E is a schematic view showing a state of a signal subjected tocomplex correlation processing of path 1;

FIG. 9F is a schematic view showing a state of a signal subjected tocomplex correlation processing of path 2;

FIG. 10 is a schematic view showing a state of a signal subjected topath combining in connection with the signal subjected to complexcorrelation processing;

FIG. 11 is a block diagram showing a configuration of a radio receivingapparatus according to Embodiment 2 of the present invention;

FIG. 12 is a block diagram showing a configuration of a radio receivingapparatus according to Embodiment 2 of the present invention; and

FIG. 13 is a block diagram showing a configuration of a radio receivingapparatus according to Embodiment 4 of the present invention.

BEST MODE FOR CARRYING OUT THE INVENTION

The following will explain the embodiments of the present invention withreference to the drawings accompanying herewith.

(Embodiment 1)

FIG. 7 is a block diagram showing a configuration of a radio receivingapparatus according to Embodiment 1 of the present invention. Inaddition, FIG. 7 is one example showing a configuration when tworeceived signals are handled and two known symbols are received withCode A and Code B in one slot by a communication partner.

In FIG. 7, a reception RF section 102 converts signals (receivedsignals) received from an antenna 101 into baseband signals using alocal signal sent from a crystal oscillator 122 to be described later,and outputs baseband signals with I-ch and Q-ch to an A/D converter 103and an A/D converter 104, respectively.

The A/D converter 103 converts the baseband signal with I-ch into adigital signal and outputs it to a searcher 105 and despreaders 106 and107. Moreover, the A/D converter 104 converts the baseband signal withQ-ch into a digital signal and outputs it to the searcher 105 and thedespreaders 106, and 107.

The searcher 105 detects despreading timing at the despreaders 106 and107, pilot timing at a channel estimating section 108, and a path timingat a RAKE combining section 109 by use of the baseband signals with I-chand Q-ch. This searcher 105 outputs the detected despread timing to thedespreaders 106 and 107, outputs the detected pilot timing to thechannel estimating section 108, and outputs the detected path timing tothe RAKE combing section 109.

The despreaders 106 and 107 provide despread processing to the basebandsignals with I-ch and Q-ch based on despread timing from the searcher105, and outputs the despread baseband signal with I-ch to a complexcorrelation section 115 and a delay section 111, and outputs thedespread baseband signal with Q-ch to the RAKE combining section 109,complex correlation section 115 and a delay section 112.

The channel estimating section 108 calculates a channel estimation valueusing the despread baseband signals from the despreaders 106 and 107based on pilot timing from the searcher 105, and outputs the calculatedchannel estimation value to the RAKE combining section 109.

The RAKE combining section 109 multiplies the despread baseband signalsfrom the despreaders 106 and 107 by an inverse characteristic of thechannel estimation value from the channel estimating section 108 basedon path timing from the searcher 105, and RAKE combines the basebandsignals multiplied by the inverse characteristic. A demodulating section110 provides demodulation processing to the baseband signals subjectedto RAKE combining.

The delay section 111 delays the baseband signal with I-ch despread bythe despreader 106, and outputs the resultant to the complex correlationsection 115. Similarly, the delay section 113 delays the baseband signalwith I-ch despread by the despreader 107, and outputs the resultant to acomplex correlation section 116. Moreover, the delay section 112 delaysthe baseband signal with Q-ch despread by the despreader 106, andoutputs the resultant to the complex correlation section 115. Likewise,the delay section 114 delays the baseband signal with Q-ch despread bythe despreader 107, and outputs the resultant to the complex correlationsection 116.

The complex correlation section 115 performs complex correlationprocessing using the baseband signal with I-ch despread by thedespreader 106 and the baseband signal with I-ch delayed by the delaysection 111. Also, the complex correlation section 115 performs complexcorrelation processing using the baseband signal with Q-ch despread bythe despreader 106 and the baseband signal with Q-ch delayed by thedelay section 112.

The complex correlation section 116 performs complex correlationprocessing using the baseband signal with I-ch despread by thedespreader 107 and the baseband signal with I-ch delayed by the delaysection 113. Also, the complex correlation section 116 performs complexcorrelation processing using the baseband signal with Q-ch despread bythe despreader 107 and the baseband signal with Q-ch delayed by thedelay section 114.

A path combining section 117 combines the signals subjected to complexcorrelation processing by the complex correlation sections 115 and 116,and outputs a combined signal to a phase estimating section 118. Thephase estimating section 118 calculates a phase rotation amount per unittime using the signals combined by the path combining section 117. Asmoothing section 119 calculates a frequency offset using the calculatedphase rotation amount per unit time. A control voltage convertingsection 120 converts the calculated frequency offset into a controlvoltage against a crystal oscillator 122. A D/A converter 121 convertsthe converted control voltage into an analog signal and outputs it tothe crystal oscillator 122. The crystal oscillator 122 is controlled bythe control voltage from the D/A converter 121 to output a local signalto the reception RF section 102.

An explanation will be next give of the operations of the radioreceiving apparatus having the aforementioned configuration withreference to FIG. 8. FIG. 8 is a schematic view showing a state of aknown symbol in a path to be received by the radio receiving apparatusaccording to Embodiment 1 of the present invention. The communicationpartner of the present radio receiving apparatus transmits a signalincluding known symbols 11 and 12, which are despread by Code A and CodeB, respectively, as illustrated in FIG. 1.

The signal transmitted from the communication partner is received viathe antenna 101 by the radio receiving apparatus shown in FIG. 7. InFIG. 7, a signal received (received signal) by the antenna 101 isconverted into baseband signals from a carrier frequency signal by thereception RF section 102. At this time, a local signal from the crystaloscillator 122 is used at the reception RF section 102. A basebandsignal with I-ch is output to the A/D converter 103 from the receptionRF section 102. Also, a baseband signal with Q-ch is output to the A/Dconverter 104 from the reception RF section 102. The baseband signalwith I-ch is converted into a digital signal by the A/D converter 103,and the digital signal is output to the searcher 105 and the despreaders106 and 107. Similarly, the baseband signal with Q-ch is converted intoa digital signal by the A/D converter 104, and the digital signal isoutput to the searcher 105 and the despreaders 106 and 107.

The searcher 105 examines the correlation between the baseband signaland Code A, which is the known code, and detects code timing with whichpower of a correlation value reaches a maximum value. Also, the searcher105 detects timing of Code B using the detected code timing. In thisway, the searcher 105 calculates despread timing at the despreaders 106and 107, pilot timing at the channel estimating section 108, and pathtiming at the RAKE combining section 109.

Timing of Code A and Code B of path 1 is output to the despreader 106and complex correlating section 115 from the searcher 105. Timing ofCode A and Code B of path 2 is output to the despreader 107 and complexcorrelating section 116 from the searcher 105. Moreover, timing of CodeA and Code B of path 1 and timing of Code A and Code B of path 2 areoutput to the channel estimating section 108 from the searcher 105.Still moreover, timing of paths 1 and 2 are output to the RAKE combiningsection 109 from the searcher 105.

The despreader 106 provides despread processing using Code A and Code Bto the baseband signal with I-ch based on timing of Code A and that ofCode B of path 1 from the searcher 105, respectively. Similarly, thedespreader 106 provides despread processing using Code A and Code B tothe baseband signal with Q-ch based on timing of Code A and that of CodeB of path 1 from the searcher 105, respectively. Moreover, thedespreader 106 provides despread processing using a predeterminedspreading code (spreading code assigned to the present radio receivingapparatus) to the baseband signals with I-ch and Q-ch.

The despreader 107 provides despread processing using Code A and Code Bto the baseband signal with I-ch based on timing of Code A and that ofCode B of path 2 from the searcher 105, respectively. Similarly, thedespreader 107 provides despread processing using Code A and Code B tothe baseband signal with Q-ch based on timing of Code A and that of CodeB of path 2 from the searcher 105, respectively. Moreover, thedespreader 107 provides despread processing using a predeterminedspreading code to the baseband signals with I-ch and Q-ch, respectively.

The baseband signals subjected to despread processing by the despreaders106 and 107 are output to the channel estimating section 108 and RAKEcombining section 109.

At the channel estimating section 108, signals corresponding to theknown symbols 11 and 12 are extracted from among the baseband signalswith I-ch and Q-ch from the despreader 106 based on timing of Code A andthat of Code B of path 1 from the searcher 105. The channel estimationvalue of path 1 is calculated using these extracted signals. Similarly,signals corresponding to the known symbols 11 and 12 are extracted fromamong the baseband signals with I-ch and Q-ch from the despreader 107based on timing of Code A and that of Code B of path 2 from the searcher105. The channel estimation value of path 2 is calculated using theseextracted signals. The channel estimation values of path 1 and path 2calculated by the channel estimating section 108 are output to the RAKEcombining section 109.

At the RAKE combining section 109, the despread baseband signals withI-ch and Q-ch from the despreader 106 are multiplied by an inversecharacteristic of the channel estimation value of path 1 from thechannel estimating section 108. The despread baseband signals with I-chand Q-ch from the despreader 107 are multiplied by an inversecharacteristic of the channel estimation value of path 2 from thechannel estimating section 108. Moreover, the despread baseband signalswith I-ch and Q-ch of path 1 multiplied by the inverse characteristic ofthe channel estimation value is RACKE combined with the despreadbaseband signals with I-ch and Q-ch of path 2 multiplied by the inversecharacteristic of the channel estimation value based on timing of path 1and timing of path 2 from the searcher 105.

The RAKE combined baseband signals with I-ch and Q-ch are output to thedemodulating section 110. At the demodulating section 110, demodulationprocessing is provided to the RAKE combined baseband signals with I-chand Q-ch, thereby obtaining received data.

On the other hand, the baseband signal with I-ch (Q-ch) subjected todespread processing by the despreader 106 is output to the complexcorrelating section 115. Also, after the baseband signal with I-ch(Q-ch) subjected to despread processing by the despreader 106 is delayedby tAB by the delay section 111 (delay section 112), the resultant isoutput to the complex correlating section 115. Here, tAB is tAB of FIG.1 and the expression of tAB=tCA/2+tgap+tCB/2 is given.

The baseband signal with I-ch (Q-ch) subjected to despread processing bythe despreader 107 is output to the complex correlating section 116.Also, after the baseband signal with I-ch (Q-ch) subjected to despreadprocessing by the despreader 107 is delayed by tAB by the delay section113 (delay section 114), the resultant is output to the complexcorrelating section 116.

The complex correlating section 115 performs complex correlationprocessing using the baseband signal with I-ch (Q-ch) subjected todespread processing by the despreader 106 and the baseband signal withI-ch (Q-ch), which is delayed by tAB by the delay section 111 (delaysection 112) and which is subjected to despread processing, based ontiming of Code A and that of Code B of path 1 from the searcher 105. Thesignals with I-ch and Q-ch of path 1 subjected to complex correlationprocessing are output to the path combining section 117.

The complex correlating section 116 performs complex correlationprocessing using the baseband signal with I-ch (Q-ch) subjected todespread processing by the despreader 106 and the baseband signal withI-ch (Q-ch), which is delayed by tAB by the delay section 111 (delaysection 112) and which is subjected to despread processing, based ontiming of Code A and that of Code B of path 2 from the searcher 105. Thesignals with I-ch and Q-ch subjected to complex correlation processingof path 2 are output to the path combining section 117.

At the path combining section 117, the signal subjected to complexcorrelation processing by the complex correlation section 115 is pathcombined with the signal subjected to complex correlation processing bythe complex correlation section 116 for each I-ch and Q-ch. The pathcombined signals with I-ch and Q-ch, which are expressed by thefollowing equation, are output to the phase estimating section 118.

$\begin{matrix}\left\{ \begin{matrix}{{{C_{est}(n)} \cdot i} = {\frac{1}{p}{\sum\limits_{P = 0}^{P - 1}{{C\left( {n,p} \right)} \cdot i}}}} \\{{{C_{est}(n)} \cdot q} = {\frac{1}{P}{\sum\limits_{P = 0}^{P - 1}{{C\left( {n,p} \right)} \cdot q}}}}\end{matrix} \right. & \left\lbrack {{Equation}\mspace{14mu} 1} \right\rbrack\end{matrix}$wherein Cest(n).i is a path-combined signal with I-ch at nth slot, andCest(n).q is a path-combined signal with Q-ch in nth slot. Also,C(n,p).i is a signal with I-ch subjected to complex correlationprocessing in nth slot and C(n,p).q is a signal with Q-ch subjected tocomplex correlation processing in nth slot.

At the phase estimating section 118, a phase rotation amount iscalculated using the signals combined by the path combining section 117.Namely, the phase rotation amount θest(n) in nth slot is expressed bythe equation given below:

$\begin{matrix}{{\theta_{est}(n)} = {\frac{1}{t_{AB}} \cdot {\tan^{- 1}\left( \frac{{C_{est}(n)} \cdot q}{{C_{est}(n)} \cdot i} \right)}}} & \left\lbrack {{Equation}\mspace{14mu} 2} \right\rbrack\end{matrix}$

The smoothing section 119 provides smooth processing to the phaserotation amount θest(n) estimated on a slot-by-slot basis by the phaseestimating section 118 based on the equation given below:φ_(est)(n)=φ_(est)(n−1)+αφ_(est)(n)[radian]  [Equation 3]wherein α is a forgetting coefficient. In addition, though thisembodiment uses a weighting average as smooth processing, a movingaverage or simple average, etc. may be used. This smooth processing cansuppress an error in the phase estimation accuracy due to noise.

Moreover, the smoothing section 119 calculates a frequency offset to becorrected by use of the phase rotation amount subjected to smoothprocessing. The frequency offset to be corrected is output to thecontrol voltage converting section 120.

The control voltage converting section 120 converts the frequency offsetto be corrected into a control voltage at the crystal oscillator 122.After this control voltage is converted into an analog signal by the D/Aconverter 121, the resultant is output to the crystal oscillator 122.The frequency offset at the crystal oscillator 122 is corrected by thecontrol voltage converted into the analog signal. By the operation ofthe aforementioned closed loop, the carrier frequency offset iscorrected at the communication partner and the present radio receivingapparatus. This makes it possible to suppress the phase rotation thatdegrades the quality of the received signal.

An explanation will be next given of the factors that make it possiblefor the radio receiving apparatus of this embodiment to estimate acorrect phase rotation amount. FIG. 10 is a schematic view showing astate of signals subjected to path combining by use of the radioreceiving apparatus according to Embodiment 1 of the present invention.

First, in FIG. 9, attention is paid to the baseband signal (hereinafterreferred to simply “baseband signal of Code A”) obtained by despreadprocessing using Code A. FIG. 9A is a schematic view showing a state ofa phase rotation amount of a baseband signal obtained by despreadprocessing using Code A of path 1. FIG. 9B is a schematic view showing astate of a phase rotation amount of a baseband signal obtained bydespread processing using Code A of path 2.

As illustrated in FIGS. 9A and 9B, the baseband signal of Code A of path1 rotates against a transmit signal by a phase variation (Δθ1fad) due tofading. While, because of a difference between path 1 and path 2 in thepropagation path, the baseband signal of Code A rotates against thebaseband signal of Code A of path 1 by a phase rotation amount (Δθp) anda phase variation (Δθ2fad) due to fading. In addition, the phaserotation amount (Δθp) is a phase rotation amount corresponding to a timedifference (tp) between path 1 and path 2.

Next, attention is paid to the baseband signal (hereinafter referred tosimply “baseband signal of Code B”) obtained by despread processingusing Code B. FIG. 9C is a schematic view showing a state of a phaserotation amount of a baseband signal obtained by despread processingusing Code B of path 1. FIG. 9D is a schematic view showing a state of aphase rotation amount of a baseband signal obtained by despreadprocessing using Code B of path 2.

As illustrated in FIGS. 9C and 9D, the baseband signal of Code B at path1 further rotates against a transmit signal of Code A of path 1 by aphase variation (ΔθAB). The baseband signal of Code B of path 2 furtherrotates against a transmit signal of Code A of path 2 by a phasevariation (ΔθAB).

The complex correlating section 115 performs complex correlationprocessing using the baseband signal of Code A and that of code B ofpath 1. FIG. 9E is a schematic view showing a state of a signalsubjected to complex correlation processing of path 1. FIG. 9F is aschematic view showing a state of a signal subjected to complexcorrelation processing of path 2.

The complex correlating section 116 performs complex correlationprocessing using the baseband signal of Code A and that of code B atpath 1 to obtain a signal subjected to complex correlation processing ofpath 1 as shown in FIG. 9E. Likewise, the complex correlating section116 performs complex correlation processing using the baseband signal ofCode A and that of code B of path 2 to obtain a signal subjected tocomplex correlation processing of path 2 as shown in FIG. 9E.

In other words, since complex correlation processing is performed usingthe baseband signals, which are not yet subjected to RAKE combining,channel estimation errors (Δθch. errA and Δθch. errB shown in FIGS. 5E,5F, 6A, and 6B), which exist in the conventional system, are notincluded in the signals subjected to complex correlation processing ofpaths 1 and 2, as is obvious from FIGS. 9E to 9F. In addition, there isa possibility that an error due to noise of the phase rotation amountwill be included in the signals subjected to complex correlationprocessing of paths 1 and 2.

After that, the aforementioned signals subjected to complex correlationprocessing of paths 1 and 2 are combined for each I-ch and Q-ch by thepath combining section 117. FIG. 10 is a schematic view showing a stateof signals subjected to path combining in connection with the signalssubjected to complex correlation processing. By path diversity effect,errors caused by noise of the phase rotation amount are reduced from thesignals subjected to path combining. As explained above, the phaseestimating section 118 estimates the phase rotation amount using thesignals subjected to path combining. As illustrated in FIG. 10, since nochannel estimation error is included in the signals subjected to pathcombining, the accuracy of phase rotation amount estimated by the phaseestimating section 118 becomes high.

In addition, FIGS. 9A, 9B, 9C, 9D, 9E, 9F and FIG. 10 shows the case inwhich the received signal exists in a first quadrant in order tosimplify the explanation. However, the present invention can be appliedto the case in which the received signal exists in any quadrant.

In this way according to this embodiment, the phase rotation amount isnot estimated using a signal obtained by combining the despread basebandsignals of the respective paths (namely, a signal obtained bymultiplying the despread baseband signal of each path by an inversecharacteristic of channel estimation value). Instead, complexcorrelation processing is performed for each path using the despreadbaseband signal, which is not yet subjected to RAKE combining (namely, abaseband signal, which is a despread signal and which is not multipliedby the inverse characteristic of channel estimation value). Then, thephase rotation amount is estimated using the signal obtained by combingthe signals subjected to complex correlation processing at therespective paths. If the phase rotation amount is thus estimated, nochannel estimation error caused by RAKE combing is included in thesignals subjected to complex correlation processing of the respectivepaths and the signals subjected to path combining, so that the phaserotation amount with good accuracy can be obtained.

Therefore, according to this embodiment, the phase rotation amount withgood accuracy can be estimated regardless of the channel estimationaccuracy even at a high speed moving time, and this makes it possible tocompensate for a frequency offset stably.

(Embodiment 2)

This embodiment explains a case in which the signals subjected to pathcombining are smoothed for each I-ch and Q-ch and the phase rotationamount is estimated using the respective smoothed signals. FIG. 11 is ablock diagram showing a configuration of a radio receiving apparatusaccording to Embodiment 2 of the present invention. In addition, somesections in the second embodiment illustrated in FIG. 11 are assignedthe same symbols as those of corresponding sections in the firstembodiment (FIG. 7) and its explanation is omitted.

The radio receiving apparatus of this embodiment is different from thatof the first embodiment in the operation after the path combiningsection 117. More specifically, the radio receiving apparatus of thisembodiment has a configuration including smoothing sections 501, 502,and a phase estimating section 503 in place of the phase estimatingsection 118 and smoothing section 119 of the radio receiving apparatusaccording to Embodiment 1.

The smoothing section 501 provides averaging processing to the signalwith I-ch subjected to path combing by the path combining section 117.The smoothing section 502 provides averaging processing to the signalwith Q-ch subjected to path combing by the path combining section 117.Here, it is assumed that a moving average in the slot is used asaveraging processing.

A signal (Cave (n).i) of nth slot subjected to the moving average by thesmoothing section 501 and a signal (Cave (n).q) of nth slot subjected tothe moving average by the smoothing section 502 can be expressed by theequation given below. In this case, k is a moving average length.

$\begin{matrix}\left\{ \begin{matrix}{{{C_{ave}(n)} \cdot i} = {\frac{1}{N}{\sum\limits_{k = 0}^{N - 1}{{C_{add}\left( {n - k} \right)} \cdot j}}}} \\{{{C_{ave}(n)} \cdot q} = {\frac{1}{N}{\sum\limits_{k = 0}^{N - 1}{{C_{add}\left( {n - k} \right)} \cdot q}}}}\end{matrix} \right. & \left\lbrack {{Equation}\mspace{14mu} 4} \right\rbrack\end{matrix}$

The phase estimating section 503 calculates the phase rotation amountusing the respective signals subjected to the moving average by thesmoothing sections 501 and 502. Namely, the phase rotation amount in thenth slot calculated by the phase estimating section 503 is expressed bythe equation given below:

$\begin{matrix}{{{\hat{\theta}}_{est}(n)} = {\frac{1}{t_{AB}} \cdot {\tan^{- 1}\left( \frac{{C_{add}(n)} \cdot q}{{C_{add}(n)} \cdot i} \right)}}} & \left\lbrack {{Equation}\mspace{14mu} 5} \right\rbrack\end{matrix}$

The phase rotation amount calculated by the phase estimating section 503is output to the control voltage converting section 120, and then thesame operation as that of Embodiment 1 is performed.

Thus, in this embodiment, the signal subjected to the path combining issmoothed for each I-ch and Q-ch, and the phase rotation amount iscalculated using the respective smoothed signals. This makes it possibleto suppress the deterioration in the phase estimation accuracy due tonoise greatly as compared with Embodiment 1 in which the calculatedphase rotation amount is smoothed.

Moreover, at a transitional time, the moving average N at the smoothingsections 501 and 502 is set to be small to make it possible to followthe change in the frequency at high-speed to estimate the frequencyoffset. Also, at a steady time, the aforementioned N is set to be largeto make it possible to adapt to the propagation path and to estimate thefrequency offset. Additionally, though this embodiment has explained thecase in which the moving average is used as smoothing processing, aweighting average or simple average may be used.

(Embodiment 3)

This embodiment explains a case in which compensation for frequencyoffset is implemented by software processing. FIG. 12 is a block diagramshowing a configuration of a radio receiving apparatus according toEmbodiment 3 of the present invention. In addition, some sections in thesecond embodiment illustrated in FIG. 12 are assigned the same symbolsas those of corresponding sections in the first embodiment (FIG. 7) andits explanation is omitted.

The radio receiving apparatus of this embodiment has a configuration inwhich a compensating section 601 is provided between the A/D converters103, 104 and the searcher 105 and the control voltage convertingsections 120, 121, and the crystal oscillator 122 of the radio receivingapparatus of Embodiment 1 are removed, and a phase vector convertingsection 602 is provided.

The phase vector converting section 602 converts the phase rotationamount (φest(n) [radian]) of the nth slot subjected to smooth processingby the smoothing section 119 into a phase rotation vector. The phaserotation vectors calculated by the phase vector converting section 602,namely, phase rotation vector (rot(n).i) of I-ch and phase rotationvector (rot(n).q) of Q-ch are expressed by the equation given below. Inthis case, K is a compensation unit in which compensation is performedone time.

$\begin{matrix}\left\{ \begin{matrix}{{{{rot}(n)} \cdot i} = {\cos\left\{ {K \cdot {{\hat{\phi}}_{est}(n)}} \right\}}} \\{{{{rot}(n)} \cdot q} = {\sin\left\{ {K \cdot {{\hat{\phi}}_{est}(n)}} \right\}}}\end{matrix} \right. & \left\lbrack {{Equation}\mspace{14mu} 6} \right\rbrack\end{matrix}$

The compensating section 601 provides compensation for frequency offsetusing the phase rotation vector of I-ch and Q-ch converted by the phasevector converting section 602 to the baseband signal with I-ch and thebaseband signal with Q-ch from the A/D converters 103 and 104. Morespecifically, the baseband signal of k sample in nth slot is s(n,k).ifrom the A/D converter 103 and the baseband signal of k sample in nthslot is s(n,k).q from the A/D converter 104. The baseband signal(sc(n,k).i) with I-ch compensated by the compensating section 601 andthe baseband signal (sc(n,k).q) with Q-ch compensated by thecompensating section 601 can be expressed by the equation given below:

$\begin{matrix}\left\{ \begin{matrix}{{{{sc}\left( {n,k} \right)} \cdot i} = {{{{s\left( {n,k} \right)} \cdot i \cdot \cos}\left\{ {k \cdot {{\hat{\phi}}_{est}(n)}} \right\}} +}} \\{{{s\left( {n,k} \right)} \cdot q \cdot \sin}\left\{ {k \cdot {{\hat{\phi}}_{est}(n)}} \right\}} \\{{{{sc}\left( {n,k} \right)} \cdot q} = {{{{- {s\left( {n,k} \right)}} \cdot i \cdot \sin}\left\{ {k \cdot {{\hat{\phi}}_{est}(n)}} \right\}} +}} \\{{{s\left( {n,k} \right)} \cdot q \cdot \cos}\left\{ {k \cdot {{\hat{\phi}}_{est}(n)}} \right\}}\end{matrix} \right. & \left\lbrack {{Equation}\mspace{14mu} 7} \right\rbrack\end{matrix}$

The baseband signal with I-ch compensated by the compensating section601 and the baseband signal with Q-ch compensated by the compensatingsection 601 are afterward subjected to processing as explained inEmbodiment 1.

In the aforementioned Embodiments 1 and 2, the carrier frequency used inthe reception RF section 102 is directly controlled such that thecrystal oscillator of the communication partner and that of the presentradio receiving apparatus become the same as each other in the accuracy,and the phase rotation due to the frequency offset is compensated.

On the other hand, in the present embodiment, the phase rotation due tothe frequency offset is compensated using digital signal processing ofthe baseband signal. According to the present embodiment, since thismakes it possible to compensate for the frequency offset by softwareprocessing such as DSP, it is possible to make the compensation accuracyof frequency offset and the correction accuracy of crystal oscillatordue to the control voltage unrelated to each other. Moreover, this canprevent deterioration in the receiving quality caused by the variationsin the parts of the crystal oscillator.

In addition, the present embodiment has explained the case in whichcompensation for frequency offset in the radio receiving apparatus ofEmbodiment 1 is implemented by software processing. However, needless tosay, compensation for frequency offset in the radio receiving apparatusof Embodiment 2 may be implemented by software processing. In this case,there can be obtained an effect that can suppress deterioration in thephase estimation accuracy in addition to the aforementioned effect.

Moreover, though the above-mentioned Embodiments 1 to 3 has explainedthe case in which the number of paths to be handled is two, the presentinvention can be applied to a case in which the number of paths to behandled is three or more and a case in which the number of paths to behandled is one. In a case where the number of paths to be handled isone, RAKE combining section 109, despreader 107, delay section 114,complex correlation section 116, and path combining section 117 may beremoved from FIGS. 7, 11, and 12. Additionally, in this case, it ispossible to estimate the phase rotation amount by the phase estimatingsection 118 using the signal subjected to complex correlation processingby the complex correlation section 115. Furthermore, after the basebandsignal subjected to the despread processing by the despreader 106 ismultiplied by the inverse characteristic of the channel estimation valuefrom the channel estimating section 108, received data can be extractedby the demodulating section 110 using the baseband signal multiplied bythe inverse characteristic.

(Embodiment 4)

This embodiment will explain a case in which space diversity is used inEmbodiments 1 to 3. Though space diversity can be applied to any one ofEmbodiments 1 to 3, a case in which space diversity is applied toEmbodiment 1 will be first explained.

FIG. 13 is a block diagram showing a configuration of a radio receivingapparatus according to Embodiment 4 of the present invention. Inaddition, some sections in the second embodiment illustrated in FIG. 13are assigned the same symbols as those of corresponding sections in thefirst embodiment (FIG. 7) and its explanation is omitted.

The radio receiving apparatus of this embodiment has a configuration inwhich a receiving system of a plurality of branches (two branches as oneexample here) and a path & branch combining section 701 for the pathcombining section 117 are added to the radio receiving apparatus ofEmbodiment 1. In addition, the receiving system is one that includesantenna 101, reception RF section 102, A/D converter 102, A/D converter103, searcher 105, despreader 106, despreader 107, delay sections 111 to114, complex correlation section 115, and complex correlation section116.

The path & branch combining section 701 performs combination of path andbranch for each I-ch and Q-ch as shown in the equation set forth below.In this case, it is assumed that the output signals of complexcorrelators 111 to 114 of nth slot, br branch, pth path are set toC(n,br,p).

$\begin{matrix}\left\{ \begin{matrix}{{{C_{est}(n)} \cdot j} = {\frac{1}{B \cdot P}{\sum\limits_{{br} = 0}^{B - 1}{\sum\limits_{p = 0}^{P - 1}{{C\left( {n,{br},p} \right)} \cdot i}}}}} \\{{{C_{est}(n)} \cdot q} = {\frac{1}{B \cdot P}{\sum\limits_{{br} = 0}^{B - 1}{\sum\limits_{P = 0}^{P - 1}{{C\left( {n,{br},p} \right)} \cdot q}}}}}\end{matrix} \right. & \left\lbrack {{Equation}\mspace{14mu} 8} \right\rbrack\end{matrix}$

The path & branch combining section 701 calculates the phase rotationamounts Cest (n).i and Cest (n).q due to the frequency offset of nthslot from the above equation.

In this way, according to this embodiment, it is possible to suppressnoise that causes deterioration in estimation accuracy of the frequencyoffset by path diversity effect and space diversity effect.Particularly, at the time of performing a burst-like reception in whicha sufficient average length required to suppress noise cannot beobtained, both path diversity effect and space diversity effect arereflected in the received signal, with the result that noise in thereceived signal can be sufficiently suppressed even if the number ofsamples of received signals is small.

Additionally, in the case where space diversity is applied to Embodiment2 (FIG. 11), the signals obtained by the path & branch combining sectionare smoothed for each I-ch and Q-ch and the phase rotation amount isestimated using the respective smoothed signals, making it possible toobtain an effect that can suppress the phase estimation accuracy due tonoise in addition to the aforementioned effect. Accordingly, it ispossible to obtain a stable reception quality as keeping the frequencyoffset accuracy more stable even when the burst-like reception isperformed.

Moreover, in the case where space diversity is applied to Embodiment 3(FIG. 12), by compensating for the phase rotation due to the frequencyoffset using digital signal processing to the baseband signal, there canbe obtained an effect that makes the compensation accuracy of frequencyoffset and the correction accuracy of crystal oscillator due to thecontrol voltage unrelated to each other in addition to theaforementioned effect. Moreover, this can prevent deterioration in thereceiving quality caused by the variations in the parts of the crystaloscillator. In this case, at the time when the signals obtained by thepath & branch combining section are smoothed for each I-ch and Q-ch andthe phase rotation amount is estimated using the respective smoothedsignals, there can be obtained an effect that can suppress the phaseestimation accuracy due to noise in addition to the aforementionedeffect.

As is obvious from the above explanation, according to the radioreceiving apparatus of the present invention, complex correlationprocessing is provided to a received signal, which is not yet subjectedto path combining, on a path-by-path basis, and a phase rotation amountis estimated from the received signal subjected to complex correlationprocessing, making it possible to correctly estimate the phase rotationamount due to a frequency offset even at a high-speed moving time.

This application is based on the Japanese Patent Application No.2000-261816 filed on Aug. 30, 2000, entire content of which is expresslyincorporated by reference herein.

INDUSTRIAL APPLICABILITY

The present invention is suitable for a radio communication apparatus.

1. A radio receiving apparatus comprising: a despreading section thatprovides despreading processing to received signals to extract in-phasecomponents and quadrature components of received signals ofpredetermined paths; a RAKE combining section that combines the in-phasecomponents and quadrature components of the despread received signals ofpredetermined paths; a complex correlation processing section thatprovides complex correlation processing to the in-phase components andquadrature components of the despread received signals of predeterminedpaths; a path combining section that combines the despread receivedsignals of predetermined paths subjected to the complex correlationprocessing per in-phase component and quadrature component to generatein-phase components and quadrature components of combined signals; and aphase rotation calculating section that calculates a phase rotationamount in the received signals using the generated in-phase componentsand quadrature components of the combined signals.
 2. The radioreceiving apparatus according to claim 1, wherein the complexcorrelation processing section provides the complex correlationprocessing to the in-phase components and quadrature components of thereceived signals of predetermined paths per predetermined path.
 3. Theradio receiving apparatus according to claim 1, wherein: the phaserotation calculating section comprises a smoothing section that providessmoothing processing to the calculated phase rotation amount; and thesmoothed phase rotation amount is used as a new phase rotation amount.4. The radio receiving apparatus according to claim 1, wherein the phaserotation calculating section comprises a smoothing section that providessmoothing processing to the generated combined signals to calculate thephase rotation amount using the combined signals subjected to thesmoothing processing.
 5. The radio receiving apparatus according toclaim 1, further comprising: a frequency offset calculating section thatcalculates a frequency offset using the phase rotation amount calculatedby the phase rotation amount calculating section; and a compensatingsection that controls a frequency of a local signal based on thecalculated frequency offset to compensate for a frequency offset in thereceived signals.
 6. The radio receiving apparatus according to claim 1,further comprising: a frequency offset calculating section thatcalculates a frequency offset using the phase rotation amount calculatedby the phase rotation amount calculating section; and a compensatingsection that provides digital calculating processing to the receivedsignals using the calculated frequency offset to compensate for thefrequency offset in the received signals.
 7. The radio receivingapparatus according to claim 1, wherein the despreading section extractsthe in-phase components and quadrature components of the receivedsignals of predetermined paths from among the received signals of aplurality of branches.
 8. A communication terminal apparatus comprisinga radio receiving apparatus, the radio receiving apparatus comprising: adespreading section that provides despreading processing to receivedsignals to extract in-phase components and quadrature components ofreceived signals of predetermined paths; a RAKE combining section thatcombines the in-phase components and quadrature components of thedespread received signals of predetermined paths; a complex correlationprocessing section that provides complex correlation processing to thein-phase components and quadrature components of the despread receivedsignals of predetermined paths; a path combining section that combinesthe despread received signals of predetermined paths subjected to thecomplex correlation processing per in-phase component and quadraturecomponent to generate in-phase components and quadrature components ofcombined signals; and a phase rotation calculating section thatcalculates a phase rotation amount in the received signals using thegenerated in-phase components and quadrature components of the combinedsignals.
 9. A radio receiving method comprising: despreading receivedsignals to extract in-phase components and quadrature components ofreceived signals of predetermined paths; RAKE combining the in-phasecomponents and quadrature components of the despread received signals ofpredetermined paths; complex correlation processing the in-phasecomponents and quadrature components of the despread received signals ofpredetermined paths; path combining the complex correlation processedsignals per in-phase component and quadrature component to generatein-phase components and quadrature components of combined signals; andcalculating a phase rotation amount in the received signals using thein-phase components and quadrature components of the generated combinedsignals.